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LTC3632 High Efficiency, High Voltage 20mA Synchronous Step-Down Converter FEATURES
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DESCRIPTION
The LTC(R)3632 is a high efficiency, high voltage step-down DC/DC converter with internal high side and synchronous power switches that draws only 12A typical DC supply current at no load while maintaining output voltage regulation. The LTC3632 can supply up to 20mA load current and features a programmable peak current limit that provides a simple method for optimizing efficiency in lower current applications. The LTC3632's combination of Burst Mode(R) operation, integrated power switches, low quiescent current, and programmable peak current limit provides high efficiency over a broad range of load currents. With its wide 4.5V to 50V input range and internal overvoltage monitor capable of protecting the part through 60V surges, the LTC3632 is a robust converter suited for regulating a wide variety of power sources. Additionally, the LTC3632 includes a precise run threshold and soft-start feature to guarantee that the power system start-up is well-controlled in any environment. The LTC3632 is available in the thermally enhanced 3mm x 3mm DFN and MS8E packages.
L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners.
Wide Input Voltage Range: Operation from 4.5V to 50V Overvoltage Lockout Provides Protection Up to 60V Internal High Side and Low Side Power Switches No Compensation Required 20mA Output Current Low Dropout Operation: 100% Duty Cycle Low Quiescent Current: 12A Wide Output Voltage Range: 0.8V to VIN 0.8V 1% Feedback Voltage Reference Adjustable Peak Current Limit Internal and External Soft-Start Precise RUN Pin Threshold with Adjustable Hysteresis Few External Components Required Low Profile (0.75mm) 3mm x 3mm DFN and Thermally Enhanced MS8E Packages
APPLICATIONS
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4mA to 20mA Current Loops Industrial Control Supplies Distributed Power Systems Portable Instruments Battery-Operated Devices Automotive Power Systems
TYPICAL APPLICATION
5V, 20mA Step-Down Converter
VIN 5V TO 50V 1F 1mH SW LTC3632 RUN VFB HYST ISET SS GND VIN VOUT 5V 10F 20mA
Efficiency and Power Loss vs Load Current
100 90 80 EFFICIENCY (%) 70 60 50 POWER LOSS 40 30 20 0.1 1 LOAD CURRENT (mA) VIN = 10V 1 VIN = 48V 10
3632 TA01b
1000 EFFICIENCY POWER LOSS (mW) 100
1.47M
280k
3632 TA01a
10
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LTC3632 ABSOLUTE MAXIMUM RATINGS (Note 1)
VIN Supply Voltage ..................................... -0.3V to 60V SW Voltage (DC) ............................-0.3V to (VIN + 0.3V) RUN Voltage .............................................. -0.3V to 60V VFB, HYST, ISET, SS Voltages......................... -0.3V to 6V Operating Junction Temperature Range (Note 2).................................................. -40C to 125C Storage Temperature Range................... -65C to 150C Lead Temperature (Soldering, 10 sec) MS8E ................................................................ 300C
PIN CONFIGURATION
TOP VIEW TOP VIEW SW VIN ISET SS 1 2 3 4 9 GND 8 7 6 5 GND HYST VFB RUN SW 1 VIN 2 ISET 3 SS 4 9 GND 8 7 6 5 GND HYST VFB RUN
MS8E PACKAGE 8-LEAD PLASTIC MSOP TJMAX = 125C, JA = 40C/W, JC = 5-10C/W EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB DD PACKAGE 8-LEAD (3mm x 3mm) PLASTIC DFN TJMAX = 125C, JA = 43C/W, JC = 3C/W EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH LTC3632EMS8E#PBF LTC3632IMS8E#PBF LTC3632EDD#PBF LTC3632IDD#PBF TAPE AND REEL LTC3632EMS8E#TRPBF LTC3632IMS8E#TRPBF LTC3632EDD#TRPBF LTC3632IDD#TRPBF PART MARKING* LTFFZ LTFFZ LFGB LFGB PACKAGE DESCRIPTION 8-Lead Plastic MSOP 8-Lead Plastic MSOP 8-Lead (3mm x 3mm) Plastic DFN 8-Lead (3mm x 3mm) Plastic DFN TEMPERATURE RANGE -40C to 125C -40C to 125C -40C to 125C -40C to 125C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
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LTC3632 ELECTRICAL CHARACTERISTICS
SYMBOL VIN UVLO PARAMETER Input Voltage Operating Range VIN Undervoltage Lockout VIN Rising VIN Falling Hysteresis VIN Rising VIN Falling Hysteresis
l l
The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are for TA = 25C (Note 2). VIN = 10V, unless otherwise noted.
CONDITIONS MIN 4.5 3.80 3.75 54 52 4.15 4.00 150 56 54 2 125 12 3
l l
TYP
MAX 50 4.50 4.35 59 57
UNITS V V V mV V V V A A A V mV nA %/V
Input Supply (VIN)
OVLO
VIN Overvoltage Lockout
IQ
DC Supply Current (Note 3) Active Mode Sleep Mode Shutdown Mode Feedback Comparator Trip Voltage Feedback Comparator Hysteresis Voltage Feedback Pin Current Feedback Voltage Line Regulation RUN Pin Threshold Voltage
VRUN = 0V VFB Rising VFB = 1V VIN = 4.5V to 50V RUN Rising RUN Falling Hysteresis RUN = 1.3V RUN < 1V, IHYST = 1mA VHYST = 1.3V VSS < 1.5V SS Pin Floating ISET Floating 500k Resistor from ISET to GND ISET Shorted to GND ISW = -10mA ISW = 10mA
l
220 22 6 0.808 7 10
Output Supply (VFB) VFB VHYST IFB VLINEREG Operation VRUN 1.17 1.06 -10 -10 4.5 40 8 1.21 1.10 110 0 0.07 0 5.5 0.75 50 25 10 5.0 2.5 60 13 1.25 1.14 10 0.1 10 6.5 V V mV nA V nA A ms mA mA mA 0.792 3 -10 0.800 5 0 0.001
IRUN VHYSTL IHYST ISS tINTSS IPEAK
RUN Pin Leakage Current Hysteresis Pin Voltage Low Hysteresis Pin Leakage Current Soft-Start Pin Pull-Up Current Internal Soft-Start Time Peak Current Trip Threshold
RON
Power Switch On-Resistance Top Switch Bottom Switch
Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC3632 is tested under pulsed load conditions such that TJ TA. LTC3632E is guaranteed to meet specifications from 0C to 85C junction temperature. Specifications over the -40C to 125C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3632I is guaranteed over the full -40C to 125C operating junction temperature range. Note that the
maximum ambient temperature consistent with these specifications is determined by specific operating conditions in conjunction with board layout, the rated package thermal impedance and other environmental factors. The junction temperature (TJ, in C) is calculated from the ambient temperature (TA, in C) and power dissipation (PD, in Watts) according to the formula: TJ = TA + (PD * JA), where JA (in C/W) is the package thermal impedance. Note 3: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information.
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LTC3632 TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Load Current, VOUT = 5V
100 95 90 EFFICIENCY (%) 85 80 75 70 65 60 55 50 0.1 VOUT = 5V FIGURE 11 CIRCUIT 1 LOAD CURRENT (mA) 10
3632 G01
TA = 25C, unless otherwise noted. Efficiency vs Load Current, VOUT = 2.5V
100 95 90 EFFICIENCY (%) 85 80 75 70 65 60 VIN = 48V VIN = 36V VOUT = 2.5V FIGURE 11 CIRCUIT 1 LOAD CURRENT (mA) 10
3632 G03
Efficiency vs Load Current, VOUT = 3.3V
100 95 90 EFFICIENCY (%) 85 80 75 70 65 60 55 50 0.1 VOUT = 3.3V FIGURE 11 CIRCUIT 1 LOAD CURRENT (mA) 10
3632 G02
VIN = 24V
VIN = 12V
VIN = 12V VIN = 24V
VIN = 12V VIN = 24V
VIN = 48V VIN = 36V
VIN = 48V VIN = 36V
55 50 0.1
Efficiency vs Input Voltage
100 95 VOUT/VOUT (%) EFFICIENCY (%) 90 85 80 ILOAD = 1mA 75 70 10 15 20 ILOAD = 5mA ILOAD = 20mA 0.50 VOUT = 5V FIGURE 11 CIRCUIT 0.40 0.30
Line Regulation
5.05 ILOAD = 20mA FIGURE 11 CIRCUIT OUTPUT VOLTAGE (V) 5.04 5.03 5.02 5.01 5.00 4.99 4.98 4.97 4.96 4.95 5 10 15 20 25 30 35 40 INPUT VOLTAGE (V) 45 50
Load Regulation
VIN = 10V VOUT = 5V FIGURE 11 CIRCUIT
0.20 0.10 0 -0.10 -0.20 -0.30 -0.40
25 30 35 40 INPUT VOLTAGE (V)
-0.50 45 50
0
10 5 15 LOAD CURRENT (mA)
20
3632 G06
3632 G04
3632 G05
Feedback Comparator Trip Voltage vs Temperature
FEEDBACK COMPARATOR TRIP VOLTAGE (V) FEEDBACK COMPARATOR HYSTERESIS (mV) 0.801 VIN = 10V 5.6 5.4 5.2 5.0 4.8 4.6
Feedback Comparator Hysteresis Voltage vs Temperature
PEAK CURRENT TRIP THRESHOLD (mA) VIN = 10V 60 50 40 30 20 10
Peak Current Trip Threshold vs Temperature and ISET
VIN = 10V ISET OPEN
0.800
RISET = 500k
0.799
ISET = GND
0.798 -40
-10
20 50 80 TEMPERATURE (C)
110
3632 G07
4.4 -40
-10
20 50 80 TEMPERATURE (C)
110
3632 G08
0 -40
-10
20 50 80 TEMPERATURE (C)
110
3632 G09
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LTC3632 TYPICAL PERFORMANCE CHARACTERISTICS
Peak Current Trip Threshold vs RISET
60 PEAK CURRENT TRIP THRESHOLD (mA) 50 40 30 20 10 0 PEAK CURRENT TRIP THRESHOLD (mA) VIN = 10V 60 50 40 30 20 10 0 0 5 10 15 20 25 30 35 40 45 50 INPUT VOLTAGE (V)
3632 G11
TA = 25C, unless otherwise noted. Quiescent Supply Current vs Input Voltage
14 12 VIN SUPPLY CURRENT (A) 10 8 6 4 2 0 5 15 25 35 INPUT VOLTAGE (V) 45
3632 G12
Peak Current Trip Threshold vs Input Voltage
VIN = 10V ISET OPEN
SLEEP
RSET = 500k
SHUTDOWN
ISET = GND
0
200
400
600 800 RISET (k)
1000
1200
3632 G10
Quiescent Supply Current vs Temperature
14 12 SLEEP 10 8 6 4 2 0 -40 SHUTDOWN SWITCH ON-RESISTANCE () VIN SUPPLY CURRENT (A) VIN = 10V 8 7 6 5 4 3 2 1 -10 20 50 80 TEMPERATURE (C) 110
3632 G13
Switch On-Resistance vs Input Voltage
8 7 SWITCH ON-RESISTANCE () 6 5 4 3 2 1 0 10 20 30 40 50
3632 G14
Switch On-Resistance vs Temperature
VIN = 10V
TOP
TOP
BOTTOM
BOTTOM
0 -40
-10
INPUT VOLTAGE (V)
20 50 80 TEMPERATURE (C)
110
3632 G15
Switch Leakage Current vs Temperature
0.30 VIN = 50V RUN COMPARATOR THRESHOLD (V) SWITCH LEAKAGE CURRENT (A) 0.25 0.20 0.15 0.10 SW = 50V 0.05 0 -40 SW = 0V 1.25 1.20 1.15 1.30
RUN Comparator Threshold Voltage vs Temperature
SWITCH VOLTAGE 20V/DIV OUTPUT VOLTAGE 100mV/DIV INDUCTOR CURRENT 50mA/DIV
Operating Waveforms
RISING
FALLING 1.10 1.05 1.00 -40
VIN = 48V VOUT = 5V ILOAD = 10mA -10 20 50 80 TEMPERATURE (C) 110
3632 G17
20s/DIV
3632 G18
-10
20 50 80 TEMPERATURE (C)
110
3632 G16
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LTC3632 TYPICAL PERFORMANCE CHARACTERISTICS
Soft-Start Waveforms
OUTPUT VOLTAGE 100mV/DIV LOAD CURRENT 20mA/DIV
Load Step Transient Response
OUTPUT VOLTAGE 2V/DIV
Short-Circuit Response
OUTPUT VOLTAGE 1V/DIV
INDUCTOR CURRENT 20mA/DIV VIN = 10V VOUT = 5V 1ms/DIV
3632 G20
CSS = 0.047F
5ms/DIV
3632 G19
VIN = 10V VOUT = 5V
200s/DIV
3632 G21
PIN FUNCTIONS
SW (Pin 1): Switch Node Connection to Inductor. This pin connects to the drains of the internal power MOSFET switches. VIN (Pin 2): Main Supply Pin. A ceramic bypass capacitor should be tied between this pin and GND (Pin 8). ISET (Pin 3): Peak Current Set Input. A resistor from this pin to ground sets the peak current trip threshold. Leave floating for the maximum peak current (50mA). Short this pin to ground for the minimum peak current (10mA). A 1A current is sourced out of this pin. SS (Pin 4): Soft-Start Control Input. A capacitor to ground at this pin sets the ramp time to full current output during start-up. A 5.5A current is sourced out of this pin. If left floating, the ramp time defaults to an internal 0.75ms soft-start. RUN (Pin 5): Run Control Input. A voltage on this pin above 1.2V enables normal operation. Forcing this pin below 0.7V shuts down the LTC3632, reducing quiescent current to approximately 3A. VFB (Pin 6): Output Voltage Feedback. Connect to an external resistive divider to divide the output voltage down for comparison to the 0.8V reference. HYST (Pin 7): Run Hysteresis Open-Drain Logic Output. This pin is pulled to ground when RUN (Pin 5) is below 1.2V. This pin can be used to adjust the RUN pin hysteresis. See Applications Information. GND (Pin 8, Exposed Pad Pin 9): Ground. The exposed pad must be soldered to the printed circuit board ground plane for optimal electrical and thermal performance.
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LTC3632 BLOCK DIAGRAM
VIN ISET 3 1A 2 C2
PEAK CURRENT COMPARATOR
RUN 5
+
1.2V
-
LOGIC AND SHOOTTHROUGH PREVENTION
HYST 7
+
REVERSE-CURRENT COMPARATOR VOLTAGE REFERENCE
FEEDBACK COMPARATOR
GND 8
GND 9
+ + -
0.800V
- +
SW 1 C1 L1 VOUT
-
5.5A
SS 4 VFB 6 R2 R1
3632 BD
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LTC3632 OPERATION (Refer to Block Diagram)
The LTC3632 is a step-down DC/DC converter with internal power switches that uses Burst Mode control, combining low quiescent current with high switching frequency, which results in high efficiency across a wide range of load currents. Burst Mode operation functions by using short "burst" cycles to ramp the inductor current through the internal power switches, followed by a sleep cycle where the power switches are off and the load current is supplied by the output capacitor. During the sleep cycle, the LTC3632 draws only 12A of supply current. At light loads, the burst cycles are a small percentage of the total cycle time which minimizes the average supply current, greatly improving efficiency. Main Control Loop The feedback comparator monitors the voltage on the VFB pin and compares it to an internal 800mV reference. If this voltage is greater than the reference, the comparator activates a sleep mode in which the power switches and current comparators are disabled, reducing the VIN pin supply current to only 12A. As the load current discharges the output capacitor, the voltage on the VFB pin decreases. When this voltage falls 5mV below the 800mV reference, the feedback comparator trips and enables burst cycles. At the beginning of the burst cycle, the internal high side power switch (P-channel MOSFET) is turned on and the inductor current begins to ramp up. The inductor current increases until either the current exceeds the peak current comparator threshold or the voltage on the VFB pin exceeds 800mV, at which time the high side power switch is turned off and the low side power switch (N-channel MOSFET) turns on. The inductor current ramps down until the reverse-current comparator trips, signaling that the current is close to zero. If the voltage on the VFB pin is still less than the 800mV reference, the high side power switch is turned on again and another cycle commences. The average current during a burst cycle will normally be greater than the average load current. For this architecture, the maximum average output current is equal to half of the peak current. The hysteretic nature of this control architecture results in a switching frequency that is a function of the input voltage, output voltage and inductor value. This behavior provides inherent short-circuit protection. If the output is shorted to ground, the inductor current will decay very slowly during a single switching cycle. Since the high side switch turns on only when the inductor current is near zero, the LTC3632 inherently switches at a lower frequency during start-up or short-circuit conditions. Start-Up and Shutdown If the voltage on the RUN pin is less than 0.7V, the LTC3632 enters a shutdown mode in which all internal circuitry is disabled, reducing the DC supply current to 3A. When the voltage on the RUN pin exceeds 1.2V, normal operation of the main control loop is enabled. The RUN pin comparator has 110mV of internal hysteresis, and therefore must fall below 1.1V to disable the main control loop. The HYST pin provides an added degree of flexibility for the RUN pin operation. This open-drain output is pulled to ground whenever the RUN comparator is not tripped, signaling that the LTC3632 is not in normal operation. In applications where the RUN pin is used to monitor the VIN voltage through an external resistive divider, the HYST pin can be used to increase the effective RUN comparator hysteresis. An internal 0.75ms soft-start function limits the ramp rate of the output voltage on start-up to prevent excessive input supply droop. If a longer ramp time and consequently less supply droop is desired, a capacitor can be placed from the SS pin to ground. The 5A current that is sourced out of this pin will create a smooth voltage ramp on the capacitor. If this ramp rate is slower than the internal 0.75ms soft-start, then the output voltage will be limited by the ramp rate
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LTC3632 OPERATION (Refer to Block Diagram)
on the SS pin instead. The internal and external soft-start functions are reset on start-up and after an undervoltage or overvoltage event on the input supply. In order to ensure a smooth start-up transition in any application, the internal soft-start also ramps the peak inductor current from 10mA during its 0.75ms ramp time to the set peak current threshold. The external ramp on the SS pin does not limit the peak inductor current during start-up; however, placing a capacitor from the ISET pin to ground does provide this capability. Peak Inductor Current Programming The offset of the peak current comparator nominally provides a peak inductor current of 50mA. This peak inductor current can be adjusted by placing a resistor from the ISET pin to ground. The 1A current sourced out of this pin through the resistor generates a voltage that is translated into an offset in the peak current comparator, which limits the peak inductor current. Input Undervoltage and Overvoltage Lockout The LTC3632 implements a protection feature which disables switching when the input voltage is not within the 4.5V to 50V operating range. If VIN falls below 4V typical (4.35V maximum), an undervoltage detector disables switching. Similarly, if VIN rises above 55V typical (53V minimum), an overvoltage detector disables switching. When switching is disabled, the LTC3632 can safely sustain input voltages up to the absolute maximum rating of 60V. Switching is enabled when the input voltage returns to the 4.5V to 50V operating range.
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LTC3632 APPLICATIONS INFORMATION
The basic LTC3632 application circuit is shown on the front page of this data sheet. External component selection is determined by the maximum load current requirement and begins with the selection of the peak current programming resistor, RISET. The inductor value L can then be determined, followed by capacitors CIN and COUT. Peak Current Resistor Selection The peak current comparator has a maximum current limit of 50mA nominally, which results in a maximum average current of 25mA. For applications that demand less current, the peak current threshold can be reduced to as little as 10mA. This lower peak current allows the use of lower value, smaller components (input capacitor, output capacitor and inductor), resulting in lower input supply ripple and a smaller overall DC/DC converter. The threshold can be easily programmed with an appropriately chosen resistor (RISET) between the ISET pin and ground. The value of resistor for a particular peak current can be computed by using Figure 1 or the following equation: RISET = IPEAK * 21 * 106 where 10mA < IPEAK < 50mA. The peak current is internally limited to be within the range of 10mA to 50mA. Shorting the ISET pin to ground programs the current limit to 10mA, and leaving it floating sets the current limit to the maximum value of 50mA. When selecting this resistor value, be aware that the maximum
1100 1000 900 800 700 RISET (k) 600 500 400 300 200 100 0 4 6 10 12 14 16 18 8 MAXIMUM LOAD CURRENT (mA) 20
average output current for this architecture is limited to half of the peak current. Therefore, be sure to select a value that sets the peak current with enough margin to provide adequate load current under all foreseeable operating conditions. Inductor Selection The inductor, input voltage, output voltage and peak current determine the switching frequency of the LTC3632. For a given input voltage, output voltage and peak current, the inductor value sets the switching frequency when the output is in regulation. A good first choice for the inductor value can be determined by the following equation: L= V VOUT * 1- OUT f * IPEAK VIN
The variation in switching frequency with input voltage and inductance is shown in the following two figures for typical values of VOUT. For lower values of IPEAK, multiply the frequency in Figure 2 and Figure 3 by 50mA/IPEAK. An additional constraint on the inductor value is the LTC3632's 100ns minimum on-time of the high side switch. Therefore, in order to keep the current in the inductor well controlled, the inductor value must be chosen so that it is larger than LMIN, which can be computed as follows: L MIN = VIN(MAX ) * tON(MIN) IPEAK(MAX )
3632 F01
Figure 1. RISET Selection
where VIN(MAX) is the maximum input supply voltage for the application, tON(MIN) is 100ns, and IPEAK(MAX) is the maximum allowed peak inductor current. Although the above equation provides the minimum inductor value, higher efficiency is generally achieved with a larger inductor value, which produces a lower switching frequency. For a given inductor type, however, as inductance is increased DC resistance (DCR) also increases. Higher DCR translates into higher copper losses and lower current rating, both of which place an upper limit on the inductance. The recommended range of inductor values for small surface mount inductors as a function of peak current is shown in Figure 4. The values in this range are a good compromise between the trade-offs discussed above. For applications
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LTC3632 APPLICATIONS INFORMATION
350 300 250 200 150 100 50 0 5 10 15 20 25 30 35 40 VIN INPUT VOLTAGE (V) 45 50 L = 1000H L = 2200H L = 470H VOUT = 5V ISET OPEN L = 220H SWITCHING FREQUENCY (kHz)
where board area is not a limiting factor, inductors with larger cores can be used, which extends the recommended range of Figure 4 to larger values. Inductor Core Selection Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of the more expensive ferrite cores. Actual core loss is independent of core size for a fixed inductor value but is very dependent of the inductance selected. As the inductance increases, core losses decrease. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates "hard," which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequently output voltage ripple. Do not allow the core to saturate! Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and do not radiate energy but generally cost more than powdered iron core inductors with similar characteristics. The choice of which style inductor to use mainly depends on the price vs size requirements and any radiated field/EMI requirements. New designs for surface mount inductors are available from Coiltronics, Coilcraft, TDK, Toko, Sumida and Vishay. CIN and COUT Selection
3632 F02
Figure 2. Switching Frequency for VOUT = 5V
250 SWITCHING FREQUENCY (kHz)
L = 220H
VOUT = 3.3V ISET OPEN
200
150
L = 470H
100 L = 1000H 50 L = 2200H
0 5 10
15 20 25 30 35 40 VIN INPUT VOLTAGE (V)
45
50
3632 F03
Figure 3. Switching Frequency for VOUT = 3.3V
10000
INDUCTOR VALUE (H)
1000
100 10 PEAK INDUCTOR CURRENT (mA)
50
3632 F04
The input capacitor, CIN, is needed to filter the trapezoidal current at the source of the top high side MOSFET. To prevent large ripple voltage, a low ESR input capacitor sized for the maximum RMS current should be used. Approximate RMS current is given by: IRMS = IOUT(MAX ) * VOUT VIN * -1 VIN VOUT
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Figure 4. Recommended Inductor Values for Maximum Efficiency
11
LTC3632 APPLICATIONS INFORMATION
This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that ripple current ratings from capacitor manufacturers are often based only on 2000 hours of life which makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. The output capacitor, COUT, filters the inductor's ripple current and stores energy to satisfy the load current when the LTC3632 is in sleep. The output voltage ripple during a burst cycle is dominated by the output capacitor equivalent series resistance (ESR) and can be estimated by the following equation: VOUT < VOUT IPEAK * ESR 160 where the lower limit of VOUT/160 is due to the 5mV feedback comparator hysteresis. The value of the output capacitor must be large enough to accept the energy stored in the inductor without a large change in output voltage. Setting this voltage step equal to 1% of the output voltage, the output capacitor must be: I COUT > 50 * L * PEAK VOUT
2
electrolytic capacitors have significantly higher ESR but can be used in cost-sensitive applications provided that consideration is given to ripple current ratings and longterm reliability. Ceramic capacitors have excellent low ESR characteristics but can have high voltage coefficient and audible piezoelectric effects. The high quality factor (Q) of ceramic capacitors in series with trace inductance can also lead to significant ringing. Using Ceramic Input and Output Capacitors Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. However, care must be taken when these capacitors are used at the input and output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN large enough to damage the part. For applications with inductive source impedance, such as a long wire, a series RC network may be required in parallel with CIN to dampen the ringing of the input supply. Figure 5 shows this circuit and the typical values required to dampen the ringing.
LIN LIN CIN CIN LTC3632 VIN R=
Typically, a capacitor that satisfies the ESR requirement is adequate to filter the inductor ripple. To avoid overheating, the output capacitor must also be sized to handle the ripple current generated by the inductor. The worst-case ripple current in the output capacitor is given by IRMS = IPEAK/2. Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Dry tantalum, special polymer, aluminum electrolytic, and ceramic capacitors are all available in surface mount packages. Special polymer capacitors offer very low ESR but have lower capacitance density than other types. Tantalum capacitors have the highest capacitance density but it is important only to use types that have been surge tested for use in switching power supplies. Aluminum
3632 F05
4 * CIN
Figure 5. Series RC to Reduce VIN Ringing
Output Voltage Programming The output voltage is set by an external resistive divider according to the following equation: VOUT = 0.8V * 1+ R1 R2
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LTC3632 APPLICATIONS INFORMATION
The resistive divider allows the VFB pin to sense a fraction of the output voltage as shown in Figure 6. Output voltage adjustment range is from 0.8V to VIN.
VOUT R1 VFB LTC3632 GND
3632 F06
R2
The RUN pin can alternatively be configured as a precise undervoltage lockout (UVLO) on the VIN supply with a resistive divider from VIN to ground. The RUN pin comparator nominally provides 10% hysteresis when used in this method; however, additional hysteresis may be added with the use of the HYST pin. The HYST pin is an opendrain output that is pulled to ground whenever the RUN comparator is not tripped. A simple resistive divider can be used as shown in Figure 8 to meet specific VIN voltage requirements.
VIN
Figure 6. Setting the Output Voltage
To minimize the no-load supply current, resistor values in the megohm range should be used; however, large resistor values should be used with caution. The feedback divider is the only load current when in shutdown. If PCB leakage current to the output node or switch node exceeds the load current, the output voltage will be pulled up. In normal operation, this is generally a minor concern since the load current is much greater than the leakage. The increase in supply current due to the feedback resistors can be calculated from: IVIN = VOUT V * OUT R1+R2 VIN
R1 RUN R2 LTC3632 HYST R3
3632 F08
Figure 8. Adjustable Undervoltage Lockout
Specific values for these UVLO thresholds can be computed from the following equations: RisingVIN UVLOThreshold = 1.21V * 1+ R1 R2 R1 R2 +R3
Run Pin with Programmable Hysteresis The LTC3632 has a low power shutdown mode controlled by the RUN pin. Pulling the RUN pin below 0.7V puts the LTC3632 into a low quiescent current shutdown mode (IQ ~ 3A). When the RUN pin is greater than 1.2V, the controller is enabled. Figure 7 shows examples of configurations for driving the RUN pin from logic.
VSUPPLY LTC3632 RUN VIN 4.7M LTC3632 RUN
FallingVIN UVLOThreshold = 1.10V * 1+
The minimum value of these thresholds is limited to the internal VIN UVLO thresholds that are shown in the Electrical Characteristics table. The current that flows through this divider will directly add to the shutdown, sleep and active current of the LTC3632, and care should be taken to minimize the impact of this current on the overall efficiency of the application circuit. Resistor values in the megohm range may be required to keep the impact on quiescent shutdown and sleep currents low. Be aware that the HYST pin cannot be allowed to exceed its absolute maximum
3632 F07
Figure 7. RUN Pin Interface to Logic
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13
LTC3632 APPLICATIONS INFORMATION
rating of 6V. To keep the voltage on the HYST pin from exceeding 6V, the following relation should be satisfied: VIN(MAX) * R3 < 6V R1+R2 +R3 A linear ramp of peak current appears as a quadratic waveform on the output voltage. For the case where the peak current is reduced by placing a resistor from ISET to ground, the peak current offset ramps as a decaying exponential with a time constant of RISET * CISET. For this case, the peak current soft-start time is approximately 3 * RISET * CISET. Unlike the SS pin, the ISET pin does not get pulled to ground during an abnormal event; however, if the ISET pin is floating (programmed to 50mA peak current), the SS and ISET pins may be tied together and connected to a capacitor to ground. For this special case, both the peak current and the reference voltage will soft-start on power-up and after fault conditions. The ramp time for this combination is CSS(ISET) * (0.8V/6A). Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% - (L1 + L2 + L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses: VIN operating current and I2R losses. The VIN operating current dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. 1. The VIN operating current comprises two components: The DC supply current as given in the electrical characteristics and the internal MOSFET gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current.
The RUN pin may also be directly tied to the VIN supply for applications that do not require the programmable undervoltage lockout feature. In this configuration, switching is enabled when VIN surpasses the internal undervoltage lockout threshold. Soft-Start The internal 0.75ms soft-start is implemented by ramping both the effective reference voltage from 0V to 0.8V and the peak current limit set by the ISET pin (10mA to 50mA). To increase the duration of the reference voltage soft-start, place a capacitor from the SS pin to ground. An internal 5A pull-up current will charge this capacitor, resulting in a soft-start ramp time given by: tSS = CSS * 0.8 V 5A
When the LTC3632 detects a fault condition (input supply undervoltage or overvoltage) or when the RUN pin falls below 1.1V, the SS pin is quickly pulled to ground and the internal soft-start timer is reset. This ensures an orderly restart when using an external soft-start capacitor. The duration of the 0.75ms internal peak current softstart may be increased by placing a capacitor from the ISET pin to ground. The peak current soft-start will ramp from 10mA to the final peak current value determined by a resistor from ISET to ground. A 1A current is sourced out of the ISET pin. With only a capacitor connected between ISET and ground, the peak current ramps linearly from 10mA to 50mA, and the peak current soft-start time can be expressed as: tSS(ISET) = CISET * 0.8 V 1A
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LTC3632 APPLICATIONS INFORMATION
2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. When switching, the average output current flowing through the inductor is "chopped" between the high side PMOS switch and the low side NMOS switch. Thus, the series resistance looking back into the switch pin is a function of the top and bottom switch RDS(ON) values and the duty cycle (DC = VOUT/VIN) as follows: RSW = (RDS(ON)TOP)DC + (RDS(ON)BOT)(1 - DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain the I2R losses, simply add RSW to RL and multiply the result by the square of the average output current: I2R Loss = IO2(RSW + RL) Other losses, including CIN and COUT ESR dissipative losses and inductor core losses, generally account for less than 2% of the total power loss. Thermal Considerations The LTC3632 does not dissipate much heat due to its high efficiency and low peak current level. Even in worst-case conditions (high ambient temperature, maximum peak current and high duty cycle), the junction temperature will exceed ambient temperature by only a few degrees. Design Example As a design example, consider using the LTC3632 in an application with the following specifications: VIN = 24V, VOUT = 3.3V, IOUT = 20mA, f = 250kHz. Furthermore, assume for this example that switching should start when VIN is greater than 12V and should stop when VIN is less than 8V. First, calculate the inductor value that gives the required switching frequency: 3.3V 3.3V * 1- L= 24V 250kHz * 50mA 220H Next, verify that this value meets the LMIN requirement. For this input voltage and peak current, the minimum inductor value is: L MIN = 24V * 100ns 48H 50mA
Therefore, the minimum inductor requirement is satisfied, and the 220H inductor value may be used. Next, CIN and COUT are selected. For this design, CIN should be size for a current rating of at least: IRMS = 20mA * 3.3V 24V * - 1 7mA RMS 24V 3.3V
Due to the low peak current of the LTC3632, decoupling the VIN supply with a 1F capacitor is adequate for most applications. COUT will be selected based on the ESR that is required to satisfy the output voltage ripple requirement. For a 50mV output ripple, the value of the output capacitor ESR can be calculated from: VOUT = 50mV 50mA * ESR A capacitor with a 1 ESR satisfies this requirement. A 10F ceramic capacitor has significantly less ESR than 1. The output voltage can now be programmed by choosing the values of R1 and R2. Choose R2 = 240k and calculate R1 as: R1= VOUT - 1 * R2 = 750k 0.8V
The undervoltage lockout requirement on VIN can be satisfied with a resistive divider from VIN to the RUN and HYST pins. Choose R1 = 2M and calculate R2 and R3 as follows: R2 = 1.21V * R1= 224k VIN(RISING) - 1.21V 1.1V VIN(FALLING) - 1.1V * R1- R2 = 90.8k
R3 =
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LTC3632 APPLICATIONS INFORMATION
Choose standard values for R2 = 226k and R3 = 91k. The ISET pin should be left open in this example to select maximum peak current (50mA). Figure 9 shows a complete schematic for this design example.
VIN 24V 1F 2M 226k 91k 220H VIN RUN SW LTC3632 ISET SS VFB HYST GND 750k 240k
3632 F09
4. Flood all unused area on all layers with copper. Flooding with copper will reduce the temperature rise of power components. You can connect the copper areas to any DC net (VIN, VOUT, GND or any other DC rail in your system).
VIN 2 1 L1 VOUT R1 COUT R2 RSET
3632 F10a
VOUT 3.3V 20mA 10F
VIN LTC3632 RUN HYST SS
SW VFB ISET GND 8, 9
5 CIN 7 4 CSS
6 3
Figure 9. 24V to 3.3V, 20mA Regulator at 250kHz
PC Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3632. Check the following in your layout: 1. Large switched currents flow in the power switches and input capacitor. The loop formed by these components should be as small as possible. A ground plane is recommended to minimize ground impedance. 2. Connect the (+) terminal of the input capacitor, CIN, as close as possible to the VIN pin. This capacitor provides the AC current into the internal power MOSFETs. 3. Keep the switching node, SW, away from all sensitive small-signal nodes. The rapid transitions on the switching node can couple to high impedance nodes, in particular VFB, and create increased output ripple.
RSET CSS VIN
L1
CIN
COUT
VOUT
R1
R2 GND
3632 F10b
VIAS TO GROUND PLANE VIAS TO INPUT SUPPLY (VIN) OUTLINE OF LOCAL GROUND PLANE
Figure 10. Layout Example
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LTC3632 TYPICAL APPLICATIONS
VIN 5V TO 50V L1 1mH VIN CIN 4.7F SW LTC3632 VFB SS GND CSS 470nF
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R1 4.2M R2 800k
COUT 100F
VOUT 5V 20mA
RUN HYST ISET
CIN: TDK C5750X7R2A475MT COUT: AVX 1812D107MAT L1: COILCRAFT LPS6235-105ML
Figure 11. High Efficiency 5V Regulator
3.3V, 20mA Regulator with Peak Current Soft-Start, Small Size
L1 470H VIN CIN 1F RUN ISET SS CSS 0.1F SW LTC3632 VFB HYST GND R1 294k R2 93.1k
3632 TA03a
Soft-Start Waveforms
VIN 4.5V TO 24V
COUT 10F
VOUT 3.3V 20mA
OUTPUT VOLTAGE 1V/DIV
INDUCTOR CURRENT 20mA/DIV 2ms/DIV
3632 TA03b
CIN: TDK C3216X7R1E105KT COUT: AVX 08056D106KAT2A L1: MURATA LQH43CN471K03
Positive-to-Negative Converter
L1 1mH CIN 1F LTC3632 RUN ISET VFB R2 71.5k VOUT -12V
3632 TA04a
Maximum Load Current vs Input Voltage
20 MAXIMUM LOAD CURRENT (mA) ISET OPEN VOUT = -3V VOUT = -5V 15
VIN 4.5V TO 38V
VIN
SW R1 1M COUT 10F
SS HYST GND
VOUT = -12V 10
CIN: TDK C3225X7R1H105KT COUT: MURATA GRM32DR71C106KA01 L1: TYCO/COEV DQ6545-102M
5 5 10
15 20 25 30 35 40 VIN INPUT VOLTAGE (V)
45
50
3632 TA04b
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17
LTC3632 TYPICAL APPLICATIONS
Small Size, Limited Peak Current, 4mA Regulator
VIN 7V TO 50V L1 2.2mH VIN CIN 1F R3 470k R4 100k R5 33k RUN SW LTC3632 VFB R1 470k R2 88.7k
3632 TA05a
COUT 10F
VOUT 5V 4mA
HYST SS ISET GND
CIN: TDK C3225X7R1H105KT COUT: AVX 08056D106KAT2A L1: MURATA LQH43NN222K03
High Efficiency 15V, 4mA Regulator
L1 10mH VIN CIN 1F SW LTC3632 VFB HYST GND R1 3M R2 169k
3642 TA07a
Efficiency vs Load Current
95 VOUT 15V 4mA EFFICIENCY (%) 90 85 80 75 70 65 60 55 50 0.1 1 LOAD CURRENT (mA) 4
3632 TA07b
VIN 15V TO 50V
VIN = 24V
COUT 4.7F
RUN ISET SS
VIN = 36V VIN = 48V
CIN: AVX 18125C105KAT2A COUT: TDK C3216X7R1E475KT L1: COILCRAFT LPS6235-106ML
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18
LTC3632 PACKAGE DESCRIPTION
DD Package 8-Lead Plastic DFN (3mm x 3mm)
(Reference LTC DWG # 05-08-1698 Rev C)
0.70 0.05
3.5 0.05 1.65 0.05 2.10 0.05 (2 SIDES) PACKAGE OUTLINE 0.25 0.05 0.50 BSC 2.38 0.05 RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED R = 0.125 TYP 5 0.40 8 0.10
3.00 0.10 (4 SIDES) PIN 1 TOP MARK (NOTE 6)
1.65 0.10 (2 SIDES)
(DD8) DFN 0509 REV C
0.200 REF
0.75 0.05
0.25
4 0.05 2.38 0.10
1 0.50 BSC
0.00 - 0.05
BOTTOM VIEW--EXPOSED PAD
NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-1) 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON TOP AND BOTTOM OF PACKAGE
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LTC3632 PACKAGE DESCRIPTION
MS8E Package 8-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1662 Rev F)
BOTTOM VIEW OF EXPOSED PAD OPTION 1 1.88 (.074) 1.68 (.066) 0.29 REF
1.88 0.102 (.074 .004)
0.889 (.035
0.127 .005)
0.05 REF 5.23 (.206) MIN 1.68 0.102 3.20 - 3.45 (.066 .004) (.126 - .136) 8 DETAIL "B" CORNER TAIL IS PART OF DETAIL "B" THE LEADFRAME FEATURE. FOR REFERENCE ONLY NO MEASUREMENT PURPOSE 0.52 (.0205) REF
0.42 0.038 (.0165 .0015) TYP
0.65 (.0256) BSC
3.00 0.102 (.118 .004) (NOTE 3)
8
7 65
RECOMMENDED SOLDER PAD LAYOUT
DETAIL "A" 0 - 6 TYP 4.90 0.152 (.193 .006) 3.00 0.102 (.118 .004) (NOTE 4)
0.254 (.010) GAUGE PLANE
1 0.53 0.152 (.021 .006) DETAIL "A" 0.18 (.007) SEATING PLANE 0.22 - 0.38 (.009 - .015) TYP 1.10 (.043) MAX
23
4 0.86 (.034) REF
NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 6. EXPOSED PAD DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH ON E-PAD SHALL NOT EXCEED 0.254mm (.010") PER SIDE.
0.65 (.0256) BSC
0.1016 (.004
0.0508 .002)
MSOP (MS8E) 0210 REV F
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LTC3632 REVISION HISTORY
REV B DATE 6/10 DESCRIPTION Text updates in Description Updates to Electrical Characteristics Updates to graphs G08, G09, G17, G18, G19 Updated description for Pins 8 and 9 in Pin Functions Text updates in Operation section Text updates in Applications Information section Figure 10 graphic added Asterisk and related text added to Typical Application Related Parts updated
(Revision history begins at Rev B)
PAGE NUMBER 1 3 4, 5, 6 6 8, 9 13 16 22 22
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
21
LTC3632 TYPICAL APPLICATION
5V, 20mA Regulator for Automotive Applications
VBATT 4.5V TO 50V TRANSIENTS UP TO 60V L1 1mH VIN CIN 1F RUN ISET SS SW LTC3632 VFB HYST GND R1 470k R2 88.7k
3632 TA06a
COUT 10F
VOUT* 5V 20mA
CIN: TDK C3225X7R2A105M COUT: KEMET C1210C106K4RAC L1: COILTRONICS DRA73-102-R
*VOUT = VBATT FOR VBATT < 5V
RELATED PARTS
PART NUMBER LTC3631/LTC3631-3.3/ LTC3631-5 LTC3642/LTC3642-3.3/ LTC3642-5 LTC1474 LT1934/LT1934-1 LT1939 LT3437 LT3470 LT3685 DESCRIPTION COMMENTS 45V, 100mA Synchronous Micropower Step-Down DC/DC Converter VIN: 4.5V to 45V (60VMAX), VOUT(MIN) = 0.8V, IQ = 12A, ISD = 3A, 3mm x 3mm DFN8, MSOP8E 45V, 50mA Synchronous Micropower Step-Down DC/DC Converter VIN: 4.5V to 45V (60VMAX), VOUT(MIN) = 0.8V, IQ = 12A, ISD = 3A, 3mm x 3mm DFN8, MSOP8E VIN: 3V to 18V, VOUT(MIN) = 1.2V, IQ = 10A, ISD = 6A, 18V, 250mA (IOUT), High Efficiency Step-Down DC/DC Converter MSOP8 VIN: 3.2V to 34V, VOUT(MIN) = 1.25V, IQ = 12A, ISD < 1A, 36V, 250mA (IOUT), Micropower Step-Down DC/DC Converter with Burst Mode Operation ThinSOTTM Package 25V, 2A, 2.5MHz High Efficiency DC/DC Converter and LDO Controller 60V, 400mA (IOUT), Micropower Step-Down DC/DC Converter with Burst Mode Operation 40V, 250mA (IOUT), High Efficiency Step-Down DC/DC Converter with Burst Mode Operation 36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High Efficiency Step-Down DC/DC Converter VIN: 3.6V to 25V, VOUT(MIN) = 0.8V, IQ = 2.5mA, ISD < 10A, 3mm x 3mm DFN10 VIN: 3.3V to 60V, VOUT(MIN) = 1.25V, IQ = 100A, ISD < 1A, 3mm x 3mm DFN10, TSSOP16E VIN: 4V to 40V, VOUT(MIN) = 1.2V, IQ = 26A, ISD < 1A, 2mm x 3mm DFN8, ThinSOT VIN: 3.6V to 38V, VOUT(MIN) = 0.78V, IQ = 70A, ISD < 1A, 3mm x 3mm DFN10, MSOP10E
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22 Linear Technology Corporation
(408) 432-1900
LT 0610 REV B * PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
FAX: (408) 434-0507 www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2009

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